Mixer circuit

ABSTRACT

A mixer circuit ( 200, 300, 800, 900 ) for mixing a first input signal at a first frequency with a second input signal at a second frequency to an output signal at a third frequency. The mixer circuit ( 200, 300, 800, 900 ) comprises a mixing stage ( 205, 805 ) with differential input ports ( 206, 207; 820, 821 ) for the first input signal and an input port ( 211, 911 ) for the second input signal and differential output ports for the output signal, which also serve as output ports for the mixer circuit. The mixer circuit ( 200, 300, 800, 900 ) comprises a nonlinear digital to analogue converter ( 210, 810 ) which has an input port ( 211 ) which is the input port for the second input signal and an output port ( 212 ) which is connected to the input port of the mixing stage, and the digital to analogue converter has a nonlinear transfer function.

TECHNICAL FIELD

The present invention discloses an improved mixer circuit.

BACKGROUND

Mixer circuits are often used in, for example, transmitters andreceivers. In, for example, a transmitter, there will often also becomprised an amplifier, usually a power amplifier. A drawback with poweramplifiers is that they usually exhibit a transfer function which isnon-linear. In order to compensate for such non-linearities in a poweramplifier, a transmitter will therefore often comprise a“pre-distortion” circuit, i.e. a circuit which introducesnon-linearities which are the inverse of the non-linearities exhibitedby the power amplifier, so that the total transfer function of thetransmitter is linearized.

A drawback of using pre-distortion circuits is that they are usuallycostly and have quite a complicated design.

SUMMARY

It is an object of the present invention to obtain a mixer circuit whichobviates at least some of the disadvantages of prior art mixer circuits,in particular when it comes to compensating for non-linearities, eitherin a power amplifier to which the mixer circuit is connected or in themixer circuit as such.

This object is addressed by the present invention in that it discloses amixer circuit for mixing a first input signal at a first frequency witha second input signal at a second frequency to an output signal at athird frequency.

The mixer circuit comprises a mixing stage with differential input portsfor the first input signal and a first input port for the second inputsignal as well as differential output ports for the output signal, whichalso serve as output ports for the mixer circuit.

In addition, the mixer circuit also comprises a digital to analogueconverter which has an input port which is the first input port for thesecond input signal and an output port which is connected to the inputport of the mixing stage. In other words, in the mixer circuit, theinput port of the digital to analogue converter is used as input portfor the second input signal. The digital to analogue converter used inthe invention has a nonlinear transfer function.

By means of the invention, as will be shown in the following detaileddescription, a mixer circuit is obtained in which compensation fornonlinearities in the mixer stage and/or in a component such as a poweramplifier to which the mixer circuit is connected is simplified ascompared to previously known solutions. In addition, the mixer circuitalso has the added advantages of low power consumption and small chipsize as compared to previously known solutions.

In embodiments of the mixer circuit, the digital to analogue convertercomprises a first and a second group of pairs of serially coupledswitches and current sources, where the number of such pairs in thefirst group corresponds to the number of digital bits in the secondinput signal. Each switch in the first group is controlled by one of thedigital bits in the second input signal, with the switches in the secondgroup being controlled to introduce the nonlinear transfer function inthe analogue to digital converter.

In embodiments of the mixer circuit, the digital to analogue convertercomprises a first and a second group of emitter coupled pairs of bipolarjunction transistors, each of which is serially connected to a currentsource via their emitters. The number of such pairs of emitter coupledbipolar junction transistors in the first group corresponds to thenumber of digital bits in the second input signal. The base of one ofeach transistors in the pairs in the first group is controlled by one ofthe digital bits in the second input signal and the base of the othertransistor in the pair is controlled by the inverse of the same digitalbit. The bases of the transistors in the pairs in the second group arecontrolled respectively by the inverse and non-inverse of one of thedigital bits in the second input signal. The transistors in the secondgroup are controlled to introduce the nonlinear transfer function in theanalogue to digital converter.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be described in more detail in the following, withreference to the appended drawings, in which

FIG. 1 shows prior art, and

FIG. 2 shows a basic embodiment of the invention, and

FIG. 3 shows a more detailed first embodiment of the invention, and

FIG. 4 shows a current source, and

FIG. 5 shows an example of a digital to analogue converter with anon-linear transfer function, and

FIG. 6 shows a table of input data and trim data to the digital toanalogue converter of FIG. 5, and

FIG. 7 shows the transfer function of the digital to analogue converterof FIG. 5, and

FIG. 8 shows a second embodiment of the invention.

FIG. 9 shows a transmitter with the mixer circuit of FIG. 3 or 8, and

FIG. 10 shows a characteristic of a VGA for use in the transmitter ofFIG. 9, and

FIG. 11 shows a VGA for use in the transmitter of FIG. 9.

FIG. 12 shows a characteristic of a second DAC.

DETAILED DESCRIPTION

Embodiments of the present invention will be described more fullyhereinafter with reference to the accompanying drawings, in whichembodiments of the invention are shown. The invention may, however, beembodied in many different forms and should not be construed as beinglimited to the embodiments set forth herein. Like numbers in thedrawings refer to like elements throughout.

The terminology used herein is for the purpose of describing particularembodiments only, and is not intended to limit the invention.

FIG. 1 shows a prior art transmitter circuit 100. The transmittercircuit 100 comprises a mixer 110 which has two input ports, indicatedas P_(in1) and P_(in2) in FIG. 1. In the circuit 100 there is alsocomprised a digital pre-distorter 115, and a digital to analogueconverter, DAC 125. The digital pre-distorter 115 receives digitalbaseband data as its input, and feeds its output to the DAC 125, whichis connected to P_(in1) of the mixer 110. Thus, one input signal to thetransmitter circuit 100 is baseband data which has been digitallypre-distorted and then converted to analogue form.

The other input port of the mixer 110, i.e. P_(in2) is used as inputfrom a second (analogue) signal, in FIG. 1 shown as the input from aLocal Oscillator, LO 130. The mixer 110 mixes the two input signals intoan output signal at a higher frequency than the frequencies of the twoinput signals, usually, as shown in FIG. 1, into an RF signal, which isfed to a power amplifier 140.

The reason for the presence of the digital pre-distorter 115 in thetransmitter circuit 100 is to “compensate” for non-linearities in thepower amplifier 140 and/or in the mixer 110. Thus, in the digitalpre-distorter 115 the input signals are distorted “inversely” todistortions which will be introduced by the power amplifier 140 and/orthe mixer 110. However, as also pointed out previously in this document,a pre-distortion circuit such as the one 115 is usually costly and hasquite a complicated design.

FIG. 2 shows a basic embodiment of a mixer circuit 200 of the invention:the mixer circuit 200 comprises a mixing stage 205 (as opposed to acomplete mixer 110, like the one shown in the embodiment 100 in FIG. 1)and a DAC 210, which has a non-linear transfer function, i.e. a“non-linear DAC”. The non-linear DAC 210 receives digital input data atan input port 211, and converts this digital input data to analogueoutput data, which is output at an output port 212. The output port ofthe non-linear DAC 210 is connected to an input port 215 of the mixerstage 205. The mixing stage 205 also comprises differential input ports206, 207 for an LO-signal, i.e. the same LO signal is input at the twodifferential ports, but with a phase shift of 180 degrees relative toeach other.

The mixing stage 205 also comprises differential output ports,symbolically shown as one ± port 220. The differential output ports ofthe mixing stage 205 also serve as output ports of the entire mixingcircuit 200.

The non-linear DAC has a non-linear transfer function, which can eitherbe made to compensate for non-linearities in the mixing stage 205 and/orknown non-linearities in other components which are not comprised in themixer circuit 200, but to which the mixer circuit will be connected,such, as, for example, a power amplifier, a PA.

FIG. 3 shows an example of a detailed embodiment 300 of the mixercircuit 200 from FIG. 2. Components which are also present in FIG. 2have retained their reference numbers from FIG. 2.

As shown in FIG. 3, the mixing stage 205 comprises an emitter coupledpair of bipolar junction transistors indicated as 330 and 331 in FIG. 2.The base of each transistor is used as one of the differential LO-inputs206, 207. The collector of each transistor 330, 331, is used as one ofthe differential outputs 221, 222, for the RF signal which is theresulting output of the mixer circuit 300. The collectors of thetransistors 330, 331 are also connected to AC ground via respectiveresistors 332, 333 which are suitably of the same resistance. Theemitters of the transistors 330, 331 are connected at a point whichserves as input port 215 to the mixing stage 205. It should be mentionedthat the bipolar junction transistors can also be replaced by sourcecoupled FET transistors. How the gate, source and drain of such FETtransistors should be connected will however not be described in detailhere.

Turning now to the embodiment of a non-linear DAC 210 which is comprisedin the mixing circuit 300, the non linear DAC 210 comprises a number ofswitches 306-310, each of which is coupled in series with a currentsource 313-317. The “pairs” of switches and current sources are dividedinto two groups, numbered as 311 and 312 in FIG. 3. The reason for thisgrouping or division will be explained below. However, with reference toFIG. 4, an example will first be given of an embodiment 400 of a currentsource for use as the current sources 313-317.

FIG. 4 shows an embodiment 400 of a so called current mirror circuit,which comprises two emitter and base-coupled bipolar junctiontransistors 401, 402. The collector of one of the transistors, in FIG. 4transistor 401, is used as output port for an output current shown as Iin FIG. 4. The collector of the other transistor, i.e. transistor 402,is connected with a bias voltage V_(b) via a resistor 403. The collectorof transistor 402 is also connected to the base of the transistor 402.The amplitude of the current I is determined by means of the biasvoltage V_(b), resistor 403, and the size of the transistors 401 and402.

The function of the nonlinear DAC 210 of FIG. 3 will now be explained inmore detail with reference to FIG. 5, which shows an enlargement of thenon-linear DAC 210. In a linear DAC, the number of pairs of switches andcurrent sources would correspond to the maximum number of input bitswhich the DAC can handle. However, as opposed to this, the non-linearDAC 210 of FIGS. 3 and 5 comprises two groups of pairs of switches andcurrent sources, i.e. the first group 311 and the second group 312. Thefirst group 311 comprises a number of pairs of current sources andswitches which corresponds to the maximum number of bits which the DAC210 is designed to handle, and the second group 312 comprises a numberof pairs of current sources and switches which are used to introduce anon-linearity in the function of the DAC 210. Thus, the first group 311of switches and current sources can also be seen as a DAC with a lineartransfer function, and the second group 312 of switches and currentsources can be seen as a “non-linearity” component in the DAC 210.

The group 312 is used to see to it that each combination of input bitsto the group 311 has its combination of “trim data”. An example of suchcombinations of input data and trim data is shown in the table of FIG.6. It should be pointed out here that the number of pairs of switchesand current sources in each of the groups 311, 312 can naturally bevaried more or less arbitrarily, so that the amounts shown in FIGS. 3and 5, i.e. three pairs in the first group 311 and two pairs in thesecond group 312 are only examples of a principle used to obtain a DACwith a non-linear transfer function.

Turning now to FIG. 6, there is shown a table of input digital data tothe first group 311 and corresponding “trim data” to the second group312 for each combination of input digital data to the first group 311.

FIG. 7 shows the transfer function of the DAC 210 (solid line), and alsoshows the transfer function of a linear DAC (dashed line) as comparison.The linear DAC used to obtain this transfer function is the one which isobtained by using only the first group 311 of switches and currentsources.

FIG. 7 thus shows the value of the output current I of the DAC 210 as afunction of the input data to the first group of switches and currentsources 311 and the trim data to the second group 312 of switches andcurrent sources. It should be pointed out that the “trim data” can ofcourse be varied depending on the non-linearity which it is desired toobtain in the DAC 210.

Thus, the nonlinear DAC 210, as shown in FIG. 5, has two current sources316, 317, which are controlled by 2 bits trim data, which are used tocontrol the position (open/closed, i.e. I/O) of the respective switch309, 310 of the current sources 316, 317. In addition, the two currentsources 316, 317 have current values indicated in FIGS. 5 and 7 as ofI_(t)/2 and I_(t). The current values as well as the “trim bits”determine the non-linear transfer function of the DAC 210. As shown inFIG. 4, the current values are determined by the bias voltages V_(b). Inconclusion, the non-linear transfer function of the DAC 210 isdetermined by the trim bits and the bias voltages of each of the currentsources in the second group 312 of switches and current sources.

The non-linear transfer function which I is desired to give the DAC 210is determined by, for example, a calibration process, following which atable, like the one in FIG. 6, is formed. The “predistortion algorithm”of the mixer circuit of the invention is thus greatly simplified ascompared to prior art: for a given input digital data, trim data isgenerated by a “look up table” such as the one in FIG. 6. The look uptable may need to be updated due to, for example, changes in ambienttemperature, which thus gives the mixer circuit of the invention anadditional function, i.e. temperature compensation.

Regarding the exact nature of the switches 306-310 of the non-linear DAC210, these can be chosen among variety of components such as, forexample, transistors, either bipolar junction transistors or FETtransistors or more “traditional” ON/OFF-switches, such as relays.

It should also be pointed out that the examples of non-linear DACs givenhere is non-exclusive, so that, in other words, other embodiments ofnon-linear DACs are also possible to use in the invention.

In order to obtain a greater degree of cancellation of LO signal leakageat the RF (output) port or ports, as well as a reduced RF signal leakageat the LO port or ports, a double balanced topology is also proposedherein, as shown in FIG. 8. Due to the similarity between the design 800shown in FIG. 8 and the design 300 of the mixer circuit shown in FIG. 3,the mixer circuit 800 will not be described in great detail here.However, as shown in FIG. 8, the mixer circuit 800 comprises a mixingstage 805 and a non-linear DAC 810. The mixing stage 805 comprises twoinput ports 820, 821, for an LO signal, and differential output ports822, 823 for a mixed signal, i.e. here an RF signal. The mixing stagecomprises two pairs of emitter coupled bipolar junction transistors 830,831 and 830′, 831′. The emitter of transistor 830′ is connected to theemitter of transistor 830, as is the emitter of transistor 831 to theemitter of transistor 831′.

The base of transistor 831′ is connected to the base of transistor 830,and these bases are used as the input for a differential LO signal, i.e.a 180 degree phase shifted LO signal.

In the non-linear DAC 810, the switches are realized as emitter coupledpairs of bipolar junction transistors 803-804, 805-806, 807-808,809-810, 811-812.

In each such pair of transistors, one transistor is controlled by a bitof binary digital data and the other transistor is controlled by theinverse of said bit, as indicated in FIG. 8 by means of input signalsS3, S3 , etc.

The embodiments of mixer circuits described above have been devoted toachieving a DAC which has a non-linear transfer function with respect tothe amplitude of the input and output signals. If it is desired tocorrect for a component which has a non-linear transfer function withrespect to both the phase of the input and output signals, theembodiments of mixer circuits described above and shown in the drawingscan also be used. This is, for example, useful in a transmitter wherethe mixer circuit is to be connected to a power amplifier whose phasevaries with the power level of the input signal, and which also has anon-linear transfer function with respect to the phase of the input andoutput signals. Such a “phase and amplitude correcting” transmitter 900is shown in FIG. 9.

The transmitter 900 comprises a mixer circuit such as the one 300 fromFIG. 3 or the one 800 from FIG. 8, and receives a first input signalfrom a Local Oscillator 905 at an input port 912, which is adifferential input port, although not explicitly shown as such in FIG.9. In addition, the mixer circuit 300 also receives digital input dataat an input port 911. The output signals from the mixer circuit 300 areused as input signals to the power amplifier 930.

In addition, the transmitter 900 also comprises a directional coupler915, by means of which a portion of the output signal from the mixercircuit 300 is diverted to 90° phase shifter 910. The output from thephase shifter 910 is used as input to a Variable Gain Amplifier, VGA920. The gain of the VGA 920 is controlled by the amplitude of the samedigital input data as is received by the mixer circuit 300 via a secondDAC 935 with a non-linear transfer function.

The output of the VGA 920 is fed to an adder 925, where it is added tothe output signals P_(PA) from the power amplifier 930, to become theoutput signal of transmitter 900.

The characteristics of a preferred VGA 920 is shown in FIG. 10, as theoutput power, P_(out), as a function of digital input data: In order tocorrect the power amplifier's 930 phase distortion, the output of theVGA 920, indicated as P_(VGA) in FIG. 10, will have a “bend” as shown inFIG. 10: as the input power to the VGA 920 is small, P_(VGA) is aroundzero, which means that no “correction” will occur. However, as the inputpower exceeds a certain value, P_(VGA) starts to increase. In thosecases, the sum of P_(VGA) and P_(PA), i.e. the output from the adder925, will obtain an extra phase shift which will counteract the phasedistortion of the power amplifier 930. Also, the amplitude of the outputsignal from the adder 925 may vary with P_(VGA), a variation that issuitably cancelled by the amplitude predistortion shown and described inconnection with FIGS. 2-8.

FIG. 11 shows an embodiment of the VGA 920 of FIG. 10. The second DAC935 with a non-linear transfer function. As shown in FIG. 11, the VGA920 comprises two emitter coupled bipolar junction transistors 13, 14,which via their coupled emitters are connected to a current source 16.The VGA has two bias voltages, V_(bb1) and V_(bb2), at the base of eachof the transistors 13, 14. The outputs from the second (non-linear) DAC935 are connected to serve as these bias voltages, V_(bb1) and V_(bb2),at the base of each of the transistors 13, 14, as shown in FIG. 12, inwhich V_(turn on) is each transistor's “turn-on voltage”. As the basebias is larger than V_(turn on), the transistor's collector starts towork as an amplifier, and when the VGA's output power is around zero,V_(bb2) is biased below V_(turn on) and V_(bb1) is biased aboveV_(turn on). Otherwise, V_(bb2) and V_(bb1) are biased in the oppositemanner.

The input digital data and the trim data to the second (non-linear) DAC935 is used to generate the desired bias voltages to the transistors 13,14, thus controlling the output power of the VGA. Each transistor 13,14, also has a bias voltage V_(CC) at its collector, in the case of thetransistor 14 via a resistor 17. The RF input signal, i.e. the signalfrom the phase shifter 910 in FIG. 9 is connected to the base oftransistor 14, together with the output of the (nonlinear) DAC 935,although the RF input signal is connected via a capacitor 16.

In the drawings and specification, there have been disclosed exemplaryembodiments of the invention. However, many variations and modificationscan be made to these embodiments without substantially departing fromthe principles of the present invention. Accordingly, although specificterms are employed, they are used in a generic and descriptive senseonly and not for purposes of limitation.

The invention is not limited to the examples of embodiments describedabove and shown in the drawings, but may be freely varied within thescope of the appended claims.

1. A mixer circuit for mixing a first input signal at a first frequencywith a second input signal at a second frequency, wherein said firstinput signal and said second input signal comprise digital input data,to an output signal at a third frequency, the mixer circuit comprising:a mixing stage with differential input ports for the first input signal,an input port for the second input signal, and differential output portsfor the output signal, wherein said differential output ports also serveas output ports for the mixer circuit; and a digital to analog converterincluding an input port, which is the input port for the second inputsignal, and an output port, which is connected to the input port of themixing stage, wherein the digital to analog converter has a nonlineartransfer function which compensates for non-linearities in the mixingstage.
 2. The mixer circuit of claim 1, wherein the digital to analogueconverter comprises a first and a second group of pairs of seriallycoupled switches and current sources, wherein the number of such pairsin the first group corresponds to the number of digital bits in thesecond input signal, and wherein each switch in the first group iscontrolled by one of the digital bits in the second input signal, withthe switches in the second group being controlled to introduce thenonlinear transfer function in the analog to digital converter.
 3. Themixer circuit of claim 2, wherein the switches comprise bipolar junctiontransistors.
 4. The mixer circuit of claim 2, wherein the switchescomprise Field Effect Transistors.
 5. The mixer circuit of claim 1,wherein the digital to analog converter comprises a first and a secondgroup of emitter coupled pairs of bipolar junction transistors, eachserially connected to a current source via their emitters, wherein thenumber of such pairs in the first group corresponds to the number ofdigital bits in the second input signal, with the base of one of eachtransistor in the pairs in the first group being controlled by one ofthe digital bits (S₃-S₁) in the second input signal and the base of theother transistor in the pair being controlled by the inverse of the samedigital bit ( S ₃- S ₁), and the bases of the transistors in the pairsin the second group are controlled respectively by the inverse ( S_(2t)- S _(1t)) and non-inverse (S_(2t)-S_(1t)) of one of the digitalbits in the second input signal, with the transistors in the secondgroup being controlled to introduce the nonlinear transfer function inthe analog to digital converter.
 6. A transmitter comprising: a mixercircuit or mixing a first input signal at a first frequency with asecond input signal at a second frequency, wherein said first inputsignal and said second input signal comprise digital input data to anoutput signal at a third frequency, the mixer circuit comprising: amixing stage with differential input ports for the first input signal,an input port for the second input signal, and differential output portsfor the output signal, wherein said differential output ports also serveas output ports for the mixer circuit; and a digital to analog converterincluding an input port, which is the input port for the second inputsignal, and an output port, which is connected to the input port of themixing stage, wherein the digital to analog converter has a nonlineartransfer function which compensates for non-linearities in the mixingstage; a Power Amplifier arranged to receive as its input signal theoutput signal from the mixer circuit; a second digital to analogconverter with a non-linear transfer function arranged to receive thesecond signal as its input data; a Variable Gain Amplifier arranged sothat its control signal is the output signal from the second digital toanalog converter; a directional coupler arranged to divert a portion ofthe output signal from the mixer circuit via a 90° phase shifter whichis also comprised in the transmitter as the input signal to the VariableGain Amplifier; and an adder arranged to add the output signals from theVariable Gain Amplifier and the Power Amplifier.